Method and apparatus to increase efficiency in a power factor correction circuit

ABSTRACT

A power factor correction (PFC) controller includes a first integrator coupled to integrate an input current of a PFC converter. A first signal is generated in response to the first integrator to end an on time of a power switch of the PFC converter. A second integrator is coupled to integrate a difference between a constant voltage and an input voltage of the PFC converter. A second signal is generated in response to the second integrator to end an off time of the power switch of the PFC converter. A driver circuit is coupled to vary the switching frequency of the power switch of the PFC converter in response to the first and the second signals and to output a third signal to switch the power switch of the PFC converter to control the input current to be substantially proportional to the input voltage.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. application Ser. No.12/267,397, filed on Nov. 7, 2008, now pending. U.S. application Ser.No. 12/267,397 is hereby incorporated by reference.

BACKGROUND

1. Field of the Disclosure

The present invention relates generally to power factor correctioncircuits in a power supply, and more specifically, the invention relatesto increasing efficiency of a power factor correction circuit.

2. Background

Power supplies are typically used to convert alternating current (“ac”)power provided by an electrical outlet into usable direct current (“dc”)power for an electrical device. One important consideration for powersupply design is how efficiently power is delivered to the power supply.To improve power delivery efficiency a power factor correction (PFC)circuit may be used in the power supply. More specifically, a powerfactor correction circuit attempts to shape the current waveform asclosely to the shape of the voltage waveform.

Typically, PFC circuits are designed to include a power switch that iscontrolled to switch between an off state and on state in order totransform a distorted input current waveform transmitted from thedistribution line into a more ideal current waveform that resembles theshape of the input voltage waveform. More specifically, the power switchis coupled to an energy transfer element to transfer energy to theoutput of the power supply. However, during operation the PFC circuitexperiences switching losses that are created in the power switch due toparasitic capacitances. Typically, parasitic capacitance can be definedas an unwanted capacitance that exists between parts of an electricalcomponent due to their proximity to each other. Additional losses arealso realized in the energy transfer element.

BRIEF DESCRIPTION OF THE DRAWINGS

Non-limiting and non-exhaustive embodiments of the present invention aredescribed with reference to the following figures, wherein likereference numerals refer to like parts throughout the various viewsunless otherwise specified.

FIG. 1 is a functional block diagram of an example boost converterincluding an example controller in accordance with the teachings of thepresent invention;

FIG. 2 is a functional block diagram further illustrating the examplecontroller of FIG. 1 in accordance with the teachings of the presentinvention;

FIG. 3A illustrates example input waveforms associated with FIGS. 1 and2 and corresponding with a switching signal and a load signal inaccordance with the teachings of the present invention;

FIG. 3B illustrates a magnified view of one of the input waveforms inFIG. 3A and a corresponding switching waveform;

FIG. 4A illustrates an example relationship between load and averageswitching frequency in accordance with the teachings of the presentinvention;

FIG. 4B illustrates an alternate example relationship between load andaverage switching frequency in accordance with the teachings of thepresent invention;

FIG. 4C illustrates an alternate example relationship between load andaverage switching frequency in accordance with the teachings of thepresent invention;

FIG. 4D illustrates an alternate example relationship between load andaverage switching frequency in accordance with the teachings of thepresent invention;

FIG. 5A illustrates an example integrated circuit that implements acontrol technique for power factor correction (PFC) in accordance withthe teachings of the present invention;

FIG. 5B illustrates an example switching frequency adjuster inaccordance with the teachings of the present invention;

FIG. 6A illustrates an example relationship between an error voltage, anadjusted error voltage, and a load in accordance with the teachings ofthe present invention;

FIG. 6B illustrates an example graph of adjusted error voltages inaccordance with the teachings of the present invention; and

FIG. 7 is a flow diagram illustrating an example method for adjustingthe average switching frequency in response to a varying load in a PFCcircuit in accordance with the teachings of the present invention.

DETAILED DESCRIPTION

In one aspect of the present invention, methods and apparatusesdisclosed here for explanation purposes use a control technique toincrease efficiency in a power factor correction (PFC) circuit. In thefollowing description, numerous specific details are set forth in orderto provide a thorough understanding of the present invention. It will beapparent, however, to one having ordinary skill in the art that thespecific detail need not be employed to practice the present invention.Well-known methods related to the implementation have not been describedin detail in order to avoid obscuring the present invention.

Reference throughout this specification to “one embodiment,” “anembodiment,” “one example” or “an example” means that a particularfeature, structure or characteristic described in connection with theembodiment is included in at least one embodiment or example of thepresent invention. Thus, the appearances of the phrases “in oneembodiment,” “in an embodiment,” “in one example” or “in an example” invarious places throughout this specification are not necessarily allreferring to the same embodiment. The particular features, structures orcharacteristics may be combined for example into any suitablecombinations and/or sub-combinations in one or more embodiments orexamples.

As will be discussed below, various examples in accordance with theteachings of the present invention implement a control technique for apower factor correction circuit to further increase power efficiencydelivery. More specifically, the control technique adjusts the averageswitching frequency of a power switch in the PFC circuit in response toa varying load coupled at the output of the PFC circuit. In oneembodiment of the present invention, the load is representative of adc-dc converter to be coupled to the output of the PFC correctioncircuit. This concept will be explained in accordance with the Figuresdescribed below.

To illustrate, FIG. 1 is a functional block diagram of an example boostPFC converter 100 (also referred to as PFC converter) including acontroller 102 in accordance with the teachings of the presentinvention. In the example shown, PFC converter 100 is a boost powerconverter that receives an ac line current I_(G) 104 which correspondswith an ac line voltage V_(G) 106. Typically, ac line current I_(G) 104and corresponding ac line voltage V_(G) 106 are provided by anelectrical distribution system (e.g., power plant) through an electricalsocket). As shown, a bridge rectifier 108 converts ac line voltage V_(G)106 to a dc input voltage V_(IN) 110.

Referring now to FIG. 3A, example waveforms 302, 304, and 306 arerepresentative of ac line voltage V_(G) 106, dc input voltage V_(IN)110, and dc input current I_(IN) 111, respectively. As shown, an ‘ac’waveform is denoted by a waveform that reverses its polarity at certainintervals. For example, ac line voltage V_(G) 106 is represented bywaveform 302 that alternates between a positive value and a negativevalue. In comparison, a ‘dc’ waveform is denoted by a waveform that isalways the same polarity. For example, as illustrated by waveforms 304and 306, dc input voltage V_(IN) 110 and a dc input current I_(IN) 111are substantially always positive. Note that dc input voltage V_(IN) 110(i.e., waveform 304) and dc input current I_(IN) 111 (i.e., waveform306) vary in magnitude with time.

Referring back to FIG. 1, in the example shown, a filter 112 is coupledacross bridge rectifier 108 to filter high frequency noise currents fromdc input current I_(IN) 111. In one aspect of the invention, dc inputcurrent I_(IN) 111 is substantially controlled to follow the waveformshape of dc input voltage V_(IN) 110. As shown in FIG. 3A waveform 306,representative of dc input current I_(IN) 111, generally follows a shapeof waveform 304, representative of dc input voltage V_(IN) 110.

As shown in the example of FIG. 1, one end of an energy storage element,shown as an inductor L₁ 114, is coupled to controller 102 while anopposite end of inductor L₁ 114 is coupled to a power switch SW₁ 118. Inoperation, power switch SW₁ 118 is in an ‘on’ or ‘closed’ state whenswitch 118 is able to conduct current and in an ‘off’ or ‘open’ statewhen switch 118 in unable to conduct current. A switching cycle isdefined as a time period when the switch is on and a subsequent timeperiod when the switch is off. For example, a switching cycle mayinclude an on time period when switch SW₁ 118 is able to conduct,followed by an off time period when switch SW₁ 118 is unable to conduct.In another example, a switching cycle may include an off time periodwhen switch SW₁ 118 is unable to conduct, followed by an on time periodwhen switch SW₁ 118 is able to conduct. An on-time may be defined as thetime period switch SW₁ 118 is conducting during a switching cycle and anoff-time may be defined as the time period switch SW₁ 118 is notconducting during a switching cycle.

In the example of FIG. 1, an input return 120 is coupled to power switchSW₁ 118. In operation, the energy storage inductor L₁ 114 transfersenergy to an output of the power converter 100 in response to theswitching of switch SW₁ 118 in accordance with the teachings of thepresent invention. As shown in the example, a bulk capacitor 122 iscoupled to supply a substantially constant output voltage V_(OUT) 124 toa load 126. In one example, load 126 may be an input to a dc-dc powersupply. A diode D1 128 is coupled such that current from bulk capacitor122 is prevented from flowing back through inductor L₁ 114. In theexample of FIG. 1, an input voltage signal U_(VIN) 130, representativeof dc input voltage V_(IN) 110, is received by controller 102. An inputcurrent sense signal U_(IIN) 132, representative of dc input currentI_(IN) 111, is also received by controller 102. More specifically, acurrent sense 134 such as for example, a current transformer, or avoltage across a discrete resistor, a voltage across a transistor whenthe transistor is conducting, or a sense FET element couple to a powerswitch may be used to measure dc input current I_(IN) 111. In theexample of FIG. 1, an output voltage signal U_(VOUT) 136, representativeof output voltage V_(OUT) 124, is also received by controller 102. Inone example, output voltage signal U_(VOUT) 136 may be representative ofa constant reference value. In one embodiment of the present invention,sense signals U_(VIN) 130, U_(IIN) 132, and U_(VOUT) 136 may be in theform of a voltage or a current.

In operation, controller 102 outputs a switching signal U_(SW) 119 thatcontrols a switching of switch SW₁ 118 in response to the input voltagesignal U_(VIN) 130, the input current signal U_(IIN) 132, and/or theoutput voltage signal U_(VOUT) 136 in order to regulate the outputvoltage V_(OUT) 124 and control the dc input current I_(IN) 111 tofollow the waveform of dc input voltage V_(IN) 110, also referred to as‘input voltage V_(IN) 110.’ In one example, controller 102 regulatesoutput voltage V_(OUT) 124 and controls dc input current I_(IN) 111 byvarying each switching cycle of switch SW₁ 118 also referred to as avariable switching frequency control technique. In another example,controller 102 regulates output voltage V_(OUT) 124 and controls dcinput current I_(IN) 111 by maintaining a constant switching cycle ofswitch SW₁ 118 also referred to as a fixed switching frequency control.In one aspect of the present invention, controller 102 also adjusts anaverage switching frequency of the switch in response to load 124 tofurther increase efficiency of PFC converter 100. In particular, theaverage switching frequency is defined as the switching frequency overmultiple switching cycles. More specifically, controller 102 employs acontrol technique that reduces power loss by adjusting an averageswitching frequency of switch SW₁ 118 in response to load 126.

Referring now to FIG. 2, a functional block diagram of power converter100 further illustrates an example of controller 102 of FIG. 1 inaccordance with the teachings of the present invention. As shown, thecontroller 102 includes power factor correction (PFC) circuitry 202 andan switching frequency adjuster 204. According to the example of FIG. 2,PFC circuitry 202 outputs switching signal U_(SW) 119 and a load signalU_(LOAD) 206 representative of the load 126 coupled to the output of PFCconverter 100. PFC circuitry 202 receives input voltage signal U_(VIN)130, output voltage signal U_(VOUT) 136, and/or input current signalU_(IIN) 134. Switching frequency adjuster 204 receives load signalU_(LOAD) 206 and outputs a frequency adjust f_(ADJ) signal 208.

In operation, as an example, the PFC circuitry 202 regulates an outputof the power supply and controls the input current such that the inputcurrent I_(IN) 111 follows the waveform of the input voltage V_(IN) 110in response to input voltage signal U_(VIN) 130, input current signalU_(IIN) 134, and/or output voltage signal U_(VOUT) 136. Switchingfrequency adjuster 204 outputs frequency adjust signal f_(ADJ) 208 inresponse to load signal U_(LOAD) 206 and PFC circuitry 202 adjusts theaverage switching frequency of power switch SW₁ 118 in response tofrequency adjust signal f_(ADJ) 208.

As shown, controller 102, current sense 134, and switch SW₁ 118 may beincluded in an integrated circuit 210. In one example, switch SW₁ 118may be included on a same single monolithic device as controller 102. Inan alternate example, controller 102 may be included on a singlemonolithic device without switch SW₁ 118. In one example, switch SW₁ 118may be a metal oxide semiconductor field effect transistor (MOSFET). Inoperation, switch SW₁ 118 allows conduction of current from a drainterminal 212 to a source terminal 214 when the switch SW₁ 118 is on andsubstantially prevents conduction of current when the switch SW₁ 118 isoff. In another example, current sense 134 may be coupled to the switchSW₁ 118 to measure a switch current I_(SW) 216 as shown. Since switchcurrent I_(SW) 216 is substantially equal to dc input current I_(IN) 111during the on time of a switching cycle (as shown in FIG. 3B), switchcurrent I_(SW) 216 may be sensed instead of dc input current I_(IN) 111during the on time of a switching cycle. As shown, current sense 134 maysense input current I_(IN) 111 at the drain terminal 212 of power switchSW₁ 118. In an alternative embodiment, current sense 134 may sense inputcurrent I_(IN) 111 at the source terminal 214 of power switch SW₁ 118.In an alternate embodiment, switch current I_(SW) 216 may be sensed bycurrent sense 134 before drain terminal 212 or after source terminal214.

As shown, switch SW₁ 118 includes parasitic capacitance C_(P) 222. Morespecifically, parasitic capacitance can be defined as an unwantedcapacitance that exists within parts of an electrical component due totheir proximity to each other. In operation, when switch SW₁ 118switches to an off state, parasitic capacitance C_(P) 222 in PFCconverter 100 stores electrical energy. Although, it is shown that straycapacitance C_(P) 22 is across power switch SW1 118, stray capacitancemay be contributing from all components within the PFC controller 100.When switch SW₁ 118 switches to an on state, the stored electricalenergy within components of PFC converter 100 is discharged and theenergy is dissipated in power switch SW₁ 118. During operation ofcontroller 100, as the switching frequency of switch SW₁ 118 increasesand switch SW₁ 118 switches more frequently between an on state and anoff state, more energy is dissipated in switch SW₁ 118. Therefore, itmay be beneficial to minimize the switching frequency of switch SW₁ 118whenever possible. In one example, the switching frequency of switch SW₁118 may be reduced as the load 126 across the output of PFC converter100 is reduced in order to limit the power dissipation in SW₁ 118.However, when limiting the frequency of SW₁ 118, losses in energytransfer element L₁ 114 are increased. More specifically, as switchingfrequency of power switch SW₁ 118 becomes lower, increases in peakcurrent may generate more power dissipation in the inductor.Additionally, losses in the energy transformer core become larger due toexcursion of magnetic flux in the core. Typically, losses in energytransfer element L₁ 114 are much greater than the prevented losses inswitch SW₁ 118 when switching frequency is reduced in a PFC converter.However, in certain PFC converter designs, it may be beneficial to lowerthe switching frequency since the losses in the power switch are greaterthan the additional power losses incurred from the energy transferelement L₁ 114.

As shown in the depicted example, filter 112 includes, but is notlimited to, a capacitor 220 that filters high frequency noise from dcinput current I_(IN) 111. More specifically, in one example, acapacitance value of capacitor 220 is a value picked such that capacitor220 may filter out high frequency noise, but is not large enough toreduce the time varying component of dc input voltage V_(IN) 110. In analternative embodiment, integrated circuit 210 may be used in PFCconverter 100 that includes a flyback converter.

As referenced previously, FIG. 3A illustrates ac line voltage waveform302, dc input voltage waveform 304, dc input current waveform 306,switch signal U_(SW) 119, and load signal U_(LOAD) 206 according to theteachings of the present invention. The ac line voltage waveform 302 isrepresentative of ac line voltage V_(G) 106 and is substantially asinusoidal waveform. A line cycle is defined as the time intervalsbetween three consecutive zero crossings of the ac line voltage waveform302 and corresponds to a line cycle period T_(L) 310 which isrepresentative of the time it takes to complete one line cycle. Morespecifically, in the example shown, the line cycle period T_(L) 310 isdependent on a frequency of the ac line voltage V_(G) 106. For example,if the frequency of the ac line voltage V_(G) 106 increases, the linecycle period T_(L) 310 will become shorter. Conversely, if the frequencyof the ac line voltage V_(G) 106 decreases, the line cycle period T_(L)310 will become longer. According to the embodiments of the presentinvention, the line cycle period T_(L) 310 is substantially longer thana switching cycle period T_(SW) 312. To further illustrate, in oneexample the line frequency is 60 Hz which corresponds to a line cycleperiod T_(L) 310 of 16,666 microseconds, and the average switchingfrequency in segment 1 is 100 kHZ which corresponds to a switching cycleperiod T_(SW) 312 of 10 microseconds.

As shown, dc input voltage waveform 304 is representative of dc inputvoltage V_(IN) 110 and is the rectified waveform of the ac line voltagewaveform 302. In operation, bridge rectifier 108 rectifies ac linevoltage V_(G) 106, represented by ac line voltage waveform 302, togenerate dc input voltage V_(IN) 110, represented by dc input voltagewaveform 304. The dc input current waveform 306 is representative of dcinput current I_(IN) 111. As shown, the dc input current waveform 306 issuperimposed on input voltage waveform 304 to illustrate how dc inputcurrent I_(IN) 111 is controlled during the switching cycles to followdc input voltage V_(IN) 110. A magnified view 314 of the dc inputcurrent waveform 306 is shown in FIG. 3B.

As shown in FIG. 3A, an average switching frequency of switching signalU_(SW) 119 varies with the magnitude of signal U_(LOAD) 206. Accordingto the teachings of the present invention, the average switchingfrequency is adjusted in response to load 126. More specifically, theaverage switching frequency is defined as the average switchingfrequency of over at least a half line cycle or more. Therefore, theaverage switching frequency can be adjusted even when the control schemeof PFC circuitry implements a variable switching frequency. As shown insegment 1, load signal U_(LOAD) 206 is at the highest magnitude andcorresponds with switching signal U_(SW) 119 being at the highestaverage switching frequency. As shown in segment 2, load signal U_(LOAD)206 is reduced in magnitude and corresponds with switching signal U_(SW)119 having a lower average switching frequency. As shown in segment 3,the load signal U_(LOAD) 206 is further reduced in magnitude whichcorresponds with switching signal U_(SW) 119 having an even loweraverage switching frequency. As shown in FIG. 3A switching input currentwaveform 306 depicts input current I_(IN) 111 in continuous conductionmode. More specifically, continuous conduction mode is a switchingcontrol technique implemented such that the input current I_(IN) 111 isprevented from reaching zero within a switching cycle because switch SW₁118 turns on before energy in energy transfer element L₁ 114 goes tozero. It will be appreciated that since input to PFC converter 100 is anac signal, that when the input voltage is zero the ac input current willbe zero even in continuous conduction mode. A discontinuous conductionmode control technique may be implemented such that the PFC controller102 prevents input current I_(IN) 111 from going to zero during each offtime of each switching cycle. In one embodiment according to the presentinvention, as the average switching frequency is reduced, PFC controller102 may switch to a discontinuous mode of operation from a continuousconduction mode of operation. Conversely, when the average switchingfrequency is increased, input current I_(IN) 111 may go from adiscontinuous mode of operation to a continuous mode of operation.

As shown in FIG. 3B, magnified view 314 depicts a portion of inputcurrent waveform 306 of FIG. 3A in continuous conduction mode. As shown,dc input current I_(IN) 111 corresponds with switching current I_(SW)216 during the on time of the switch. In operation input current I_(IN)111 is controlled for a first switching cycle period T_(SW1) 316 inresponse to a first on time T_(ON1) 318 and a first off time T_(OFF1)320 that is determined by controller 102. As illustrated in magnifiedview 314, switching cycles T_(SW1), T_(SW2), and T_(SW3) are varied dueto a variable frequency control technique implemented by controller 102for PFC correction. Therefore, according to one embodiment of thepresent invention, controller 102 may adjust the average switchingfrequency of switch S_(W1) 118 over multiple line cycles in response tovarying loads and may also adjust the cycle by cycle switching frequencyto regulate the output voltage and control the input current I_(IN) 111to follow input voltage V_(IN) 110 for PFC.

Referring now to FIGS. 4A, 4B, 4C, and 4D, the average switchingfrequency may be adjusted in various ways in response to the load of thePFC converter 100. In FIG. 4A, the average switching frequency of switchSW₁ 118 varies linearly and continuously with load 126. In FIG. 4B, theaverage switching frequency of switch SW₁ 118 varies exponentially andcontinuously with load 126. In FIG. 4C, the average switching frequencyof switch SW₁ 118 varies linearly and discretely with load 126.According to an embodiment of the present invention and as illustratedin FIGS. 4A, 4B, and 4C, the average switching frequency is adjustedover the entire load range of the boost converter. That is, the averageswitching frequency may be at a minimum frequency in response to aminimum load condition and the average switching frequency may be at amaximum frequency in response to a maximum load condition of the boostconverter. In another embodiment, as shown in FIG. 4D the averageswitching frequency is adjusted linearly and continuously over a portionof the entire load range. In another embodiment of the presentinvention, FIGS. 4A, 4B, 4C, and 4D may be combined in any type ofcombination to determine the relationship between average switchingfrequency of switch SW₁ 118 and load 126.

Referring now to FIG. 5A, an example integrated circuit controller 500that uses a particular control technique to implement PFC and furtherincludes a switching frequency adjuster 501 to improve efficiency inaccordance with the teachings of the present invention is shown. In theexample, a power MOSFET 502 switches between an on state and an offstate to permit and prevent a flow of input current I_(IN) 503 between adrain terminal D 504 and a source terminal S 506. A voltage terminalV_(IN) 507 is coupled to receive input voltage signal V_(IN) 110. Asshown, an input voltage detector 508 outputs a current MN 512representative of an instantaneous dc input voltage of PFC converter100. In operation, a generated current signal I_(VIN2) 513 is derivedfrom input voltage detector 508. According to the teachings of theembodiment in the present invention, generated current signal I_(VIN2)513 may be representative of a peak input voltage of a half-line cycle,either/or an rms value of the input voltage, either/or an average inputvoltage over a half cycle. A feedback terminal FB 514 receives a voltageV_(VOUT) representative of an output voltage at the output of PFCconverter 100. In one example, voltage V_(VOUT) may be any constantvalue.

As shown, a reference current I_(REF) flows from a current source 520 inthe opposite direction of a scaled current I_(SVIN) which flows from acurrent source 522. More specifically, scaled current I_(SVIN) is equalto current MN multiplied by a scaling factor for signal processing. Acapacitor C_(OFF) 524 is coupled across a transistor T_(OFF) 526. Inoperation, capacitor C_(OFF) 524 charges when transistor T_(OFF) 526 isoff. More specifically, the current that charges capacitor C_(OFF) 524is the difference between reference current I_(REF) and scaled currentI_(SVIN). When transistor T_(OFF) 526 turns on, capacitor C_(OFF) 524discharges via a common return 529. A voltage comparator 528 is coupledto capacitor C_(OFF) 524 such that a negative terminal of the comparator528 is at a same potential voltage as the capacitor C_(OFF) 524. Whenthe voltage on capacitor C_(OFF) 524 equals an adjusted error voltageV′_(ERR) 530, a voltage signal V_(OFF) 532 transitions from low to highwhich results in power MOSFET 502 transitioning to an on state. In thismanner, the off time of a switching cycle for power MOSFET 502 isadjusted. In one embodiment of the present invention, capacitor C_(OFF)524 functions as an integrator that integrates a difference between aconstant voltage and an input voltage of a power converter to determinethe off time of a switching cycle.

In operation, in the example shown, an error voltage V_(ERR) 531 is anoutput of error amplifier 533. In operation, error amplifier 533compares voltage V_(VOUT) with a reference voltage V_(REF) 535 todetermine error voltage V_(ERR) 531 which is representative of theoutput voltage at the output of a power converter. According to anembodiment of the present invention voltage error signal V_(ERR) 531gives an indication of the output voltage of the power converter 100 andalso the load at the output of the controller. According to theteachings of the present invention, the error signal V_(ERR) 531 isdesigned to have a substantially slower response time in comparison tothe switching signal U_(SW) 119 (e.g., drive signal 554). For example,in one embodiment, error signal V_(ERR) 531 is an averaged valuerepresentative of an averaged magnitude of output voltage V_(OUT) 124over several line cycles such that output voltage V_(OUT) 124 isconsidered a substantially constant value when controlling the inputcurrent over a line cycle.

More specifically, in this example, error signal V_(ERR) 531 issubstantially non-responsive to ac time variances in the output voltageV_(OUT) 124 over a line cycle. It can also be assumed that error signalV_(ERR) 531 is substantially constant over multiple switching cycles. Inone example, error voltage V_(ERR) 531 may be output via a COMP terminal537 to a gain setting filter that adjusts a response time of errorvoltage V_(ERR) 531.

As shown in the example, a switching frequency adjuster 501 is coupledbetween the output of error amplifier 533 and a non inverting terminalof comparator 528. Switching frequency adjuster 501 is one possibleimplementation of switching frequency adjustor 204 of FIG. 2, while someor all of the remaining circuitry of integrated circuit controller 500is a possible implementation of PFC circuitry 202. In operation,switching frequency adjuster 501 outputs an adjusted error signalV_(ERR) 530 in response to receiving an error voltage signal V_(ERR)531. In one example according to the embodiment of the present inventionthe error voltage signal V_(ERR) 531 is modified based on the followingequation:

V′ _(ERR) =V _(C) −V _(ERR)  EQ. 1

where V_(C) is a constant value determined based on design parameters ofthe feedback loop of PFC converter 100. In accordance with the teachingsof the present invention, error voltage signal V_(ERR) 531 can beconsidered equivalent to load signal U_(LOAD) 206 shown in FIG. 2. Morespecifically, the magnitude of the error voltage signal V_(ERR) 531 isdirectly proportional to the load at the output of the power converter.Typically, the magnitude of error voltage signal V_(ERR) 531 may beinfluenced by a varying input voltage as well as a changing loadcondition, therefore a directly proportional relationship between themagnitude of the feedback signal V_(ERR) 531 and the load condition atthe output of the power converter may be difficult to establish.According to the embodiment of the present invention, input voltagedetector generates signal current I_(VIN2) which is representative of anaverage input voltage value and is multiplied with current source 534 tooffset the effects of input voltage on the magnitude of error voltagesignal V_(ERR) 531. In other words, current I_(VIN2) is multiplied tocurrent source 534 in a feed forward system such that the magnitude oferror voltage signal V_(ERR) 531 is independent of input voltage V_(IN)111 and may be representative of the load 126 coupled to the output ofPFC converter 100. In one embodiment according to the present inventionadjusted error signal V′ERR 530 can be considered the equivalent offrequency adjust signal f_(ADJ) 208 in FIG. 2.

In another example embodiment according to the present invention,switching frequency adjuster 501 is coupled between error amplifier 533and the positive input terminal of voltage comparator 544. In operationof this example the adjusted error signal V′_(ERR) 530 is received byvoltage comparator 544.

As shown in the example of FIG. 5A, a current source 534 outputs ascaled current I_(SIIN) that is representative of a sensed input currentI_(S) 538 multiplied by a scaling factor for signal processing. Acapacitor C_(ON) 540 is coupled across a transistor TON 542. Inoperation, scaled current I_(SIIN) charges capacitor C_(ON) 540 whentransistor TON 542 is off. When transistor TON 542 is on, capacitorC_(ON) 540 discharges via common return 529. Voltage comparator 544 iscoupled to capacitor C_(ON) 540 such that a negative input of thecomparator 544 is at the same potential voltage as the capacitor C_(ON)540. When the voltage on capacitor C_(ON) 540 equals error voltageV_(ERR) 531, a voltage signal VON 546 at an output of comparator 544transitions from a low signal to a high signal, which results in settingpower MOSFET 502 to an off state. In this manner, the on time of aswitching cycle for power MOSFET 502 is controlled. In one aspect of thepresent invention, the capacitor C_(ON) 540 functions as an integratorthat integrates an input current of a power converter to determine theon time of a switching cycle.

As shown in the example of FIG. 5A, a first input of an OR gate 548 iscoupled to the output of comparator 544 and a second input of OR gate548 is coupled to an output of an AND gate 550. In operation, OR gate548 outputs a high signal to reset (R) of latch 552 when the voltagesignal VON 546 transitions high or an over current protection (OCP)signal 553 transitions high. In operation, when reset input R of latch552 receives a high signal, output Q is set high and complementaryoutput Q is set low. Conversely, when voltage signal V_(OFF) 532transitions high, input S of latch 552 sets output Q low andcomplementary output Q is set high. In this manner, complementary outputQ outputs a drive signal DRIVE 554 that controls a switching of powerMOSFET 502. An amplifier 556, amplifies drive signal DRIVE 554 in orderto supply adequate current to charge and discharge the gate of powerMOSFET 502 to control the switching of power MOSFET 502.

As shown in the example, a current limit comparator 558 compares sensedinput current I_(S) 538 with a current limit reference I_(LIM) 559. Inone example, the output of current limit comparator 558 goes high whenthe sensed input current I_(S) 538 reaches the current limit referenceI_(LIM) 559. More specifically, in this example, sensed input currentI_(S) 538 is a portion of input current IIN 503. In one example, sensedinput current I_(S) 538 is representative of input current IIN 503 inaccordance with the teachings of present invention. In one example,drive signal 554 is delayed by leading edge blanking (LEB) circuit 562before being applied to the input of AND gate 550 to prevent the overcurrent protection signal 553 from indicating a false current limitcondition when power MOSFET 502 momentarily discharges stray capacitanceas it turns on. More specifically, over current protection signal 553indicates when the current in power MOSFET 502 has reached the currentlimit reference I_(LIM) 559, to prevent damage to the power MOSFET 502and/or any other internal components of integrated circuit 500.

As is discussed above the teachings of the present invention allow apower converter to employ a control technique to shape an input currentwaveform of the power converter. In addition average frequency adjustcircuit 501 is included to adjust the average switching frequency ofMOSFET 502 along all load ranges to limit losses in the controller. Inthe examples discussed, the PFC controller controls the input currentwaveform to follow the shape of an input voltage waveform by varying anon time and an off time of a power switch in the power converter. Morespecifically, the input current is controlled to be directlyproportional to the input voltage over each half line cycle. However,when the input current is averaged out over multiple half line cyclesinput current I_(IN) 111 is no longer proportional to the input voltageV_(IN) 110. More specifically, the control technique forces the on timeof the power switch to be inversely proportional to a rectifiedtime-varying input voltage V_(IN)(t) by setting a constant volt-secondsfor the off time. The off-time is controlled to be a constant productof:

(V _(OUT) −V _(IN))×T _(OFF)  EQ. 2

In particular, integrating the quantity:

V _(OUT) −V _(IN)  EQ. 3

during the off time allows for a constant volt-seconds to be set duringthe off time. By setting the off time to have a constant volt-seconds,the on-time volt-seconds is forced to be substantially constant over afew switching cycles in order to maintain a volt-second balance thatsatisfies the properties of a boost inductor. A balance of volt-secondson the boost inductor allows the on-time to be substantially inverselyproportional to the input voltage. This relationship of on-time to inputvoltage sets up a convenient and simple means for controlling the inputcurrent as a function of the rectified time varying input voltageV_(IN)(t) which is representative of the input line voltage. If theinput current is sensed by integrating the input current during the ontime, the on time can be terminated by reaching a constant integralvalue of:

$\begin{matrix}{\int_{T\; 1}^{T\; 2}{I_{INPUT}\ {t}}} & {{EQ}.\mspace{14mu} 4}\end{matrix}$

where the duration from T1 to T2 is the on time as determined by thesubstantially constant feedback signal over a few switching cycles. Thiswill cause average input current over a switching cycle to besubstantially proportional to the input voltage.

Referring now to FIG. 5B, an example switching frequency adjuster 501includes a current mirror 560 in accordance with the teachings of thepresent invention. As shown, current mirror 560 further includes a gateof a first transistor T1 562 coupled to a gate of a second transistor T2564. In operation, current mirror 560 receives first current I₁ tocontrol a second current I₂ through second transistor T2 564. Morespecifically, error voltage signal V_(ERR) 531 is converted to firstcurrent I₁ by a first resistor R₁ 566 and second current I₂ isproportional to first current I₁ due to the current mirrorconfiguration. In other words, as first current I₁ increases the secondcurrent I₂ will increase proportionately. A second resistor R₂ 568converts the constant voltage V_(C), discussed in FIG. 5A, to secondcurrent I₂. Adjusted error voltage V′_(ERR) 530 is determined. In anexample operation, as first current I₁ increases due to an increase inerror voltage signal V′_(ERR) 530, the second current I₂ increases inproportion. As second current I₂ increases, a voltage drop across secondresistor R₂ 568 increases thus reducing the magnitude of the adjusterror voltage signal V′_(ERR) 530. According to the embodiment in FIG.5A, since adjusted error signal V′_(ERR) 530 functions as a threshold todetermine the off time, as shown in the example control circuit of FIG.5A, the adjusted error voltage signal V′_(ERR) 530 adjusts the averageswitching frequency of the switch by varying the off time of powerswitch SW₁ 118 during a switching cycle

Referring now to FIG. 6A, an example graph 600 further illustrates thefunctional relationship of adjusted error voltage signal V′ERR 530 as afunction of error voltage signal V_(ERR) 531 implemented by averagefrequency adjuster 501 according to an embodiment of the presentinvention. In one example, error voltage signal V_(ERR) 531 isrepresentative of load signal U_(LOAD) 206 and adjusted error voltagesignal V′_(ERR) 530 is representative of frequency adjust signal f_(ADJ)208. As shown, the magnitude of voltage error signal V_(ERR) 531increases as load increases. Conversely, adjusted error voltage signalV′_(ERR) 530 decreases as load increases. As stated above, in oneexample, a relationship between error voltage V_(ERR) 531, adjustederror voltage V′_(ERR) 530, and a constant voltage V_(C) is describedbelow:

V′ _(ERR) =V _(C) V _(ERR)  EQ. 5

where V_(C) is a constant voltage that is picked in accordance with thedesign parameters of the feedback loop in the PFC converter 100. Inalternate embodiments according to the teachings of the presentinvention, graph 600 may resemble other functional relationships or acombination of the other functional relationships between averageswitching frequency and load as illustrated in FIGS. 4A, 4B, 4C, and 4D.

Referring now to FIG. 6B, a graph 650 illustrates a first adjusted errorvoltage V′_(ERR1) and a second adjusted voltage V′_(ERR2). As shown,adjusted error voltage V′_(ERR) 530 has an inverse relationship with theload, therefore when the adjusted error voltage V′_(ERR) 530 isincreased from V′_(ERR1) to V′_(ERR2) it is representative of adecreasing load at the output of PFC converter 100.

Referring back to FIG. 5, when adjusted error voltage threshold V′_(ERR)530 increases it will take a longer time for the voltage acrosscapacitor C_(OFF) 524 to meet the increased adjusted threshold andtherefore delay the time the MOSFET 502 is turned on again, thuslengthening the off time of the switching cycle. In this manner theaverage switching frequency is reduced when load 126 at the output ofPFC converter 100 is reduced. Due to the control technique implementedin integrated controller 500 the on time of each switching cycle is alsostretched so as to maintain to the same duty ratio. Conversely, theaverage switching frequency can be increased when load 126 increases theadjusted voltage reference coupled to positive terminal of voltagecomparator 528 thus reducing the time it takes to charge C_(ON) 540 andleading to a shorter off time.

FIG. 7 is a flow diagram illustrating an example method for adjustingthe switching frequency in response to a varying load conditionaccording to the teachings of the present invention. In a process block710, switch SW₁ 118 is switched to regulate an output voltage and toimplement power factor correction such that the input current I_(IN) 111is directly proportional to the input current V_(IN) 110. In a nextprocess block 715, PFC circuitry 212 outputs a load signal U_(LOAD) 206representative of load 126 to an average frequency adjuster 204. In anext process block 720, switching frequency adjuster outputs frequencyadjust signal f_(ADJ) 208 to PFC circuitry to adjust the averageswitching frequency in response to the load signal U_(LOAD) 206. In anext process block 725, PFC circuitry adjusts switching signal U_(SW)119 such that the average switching frequency of power switch SW₁ 1118is adjusted in response to load 126 coupled to output of PFC converter100. After execution of decision block 725, the process returns back toprocess block 710.

The above description of illustrated examples of the present invention,including what is described in the Abstract, are not intended to beexhaustive or to be limitation to the precise forms disclosed. Whilespecific embodiments of, and examples for, the invention are describedherein for illustrative purposes, various equivalent modifications arepossible without departing from the broader spirit and scope of thepresent invention.

These modifications can be made to examples of the invention in light ofthe above detailed description. The terms used in the following claimsshould not be construed to limit the invention to the specificembodiments disclosed in the specification and the claims. Rather, thescope is to be determined entirely by the following claims, which are tobe construed in accordance with established doctrines of claiminterpretation. The present specification and figures are accordingly tobe regarded as illustrative rather than restrictive.

1. A power factor correction (PFC) controller, comprising: a firstintegrator coupled to integrate an input current of a PFC converter,wherein a first signal is generated in response to the first integratorto end an on time of a power switch of the PFC converter; a secondintegrator coupled to integrate a difference between a constant voltageand an input voltage of the PFC converter, wherein a second signal isgenerated in response to the second integrator to end an off time of thepower switch of the PFC converter; and a driver circuit coupled to varythe switching frequency of the power switch of the PFC converter inresponse to the first and the second signals and to output a thirdsignal to switch the power switch of the PFC converter to control theinput current to be substantially proportional to the input voltage. 2.The PFC controller of claim 1 further comprising a frequency adjustcircuit coupled to receive an error voltage signal representative of anoutput of the PFC converter, wherein the frequency adjust circuit isfurther coupled to output an adjusted error signal to adjust the end ofthe off time of the power switch of the PFC converter in response to theerror voltage signal.
 3. The PFC controller of claim 2 wherein thedriver circuit is coupled to regulate an output voltage of the PFCconverter.
 4. The PFC controller of claim 1 wherein the power switch ofthe PFC converter comprises a metal oxide field effect transistor(MOSFET).
 5. The PFC controller of claim 2 further comprising an erroramplifier coupled to generate the error voltage signal in response to afeedback signal representative of the output of the PFC converter and areference voltage.
 6. The PFC controller of claim 1 wherein the firstintegrator includes a first capacitor coupled to be charged with ascaled current when the power switch is on, wherein the scaled currentis representative of the input current.
 7. The PFC controller of claim 6wherein a scaling factor of the scaled current is representative of apeak input voltage of a half-line cycle.
 8. The PFC controller of claim6 wherein a scaling factor of the scaled current is representative of anrms value of the input voltage.
 9. The PFC controller of claim 6 whereina scaling factor of the scaled current is representative of an averageinput voltage over a half cycle.
 10. The PFC controller of claim 6further comprising a first comparator coupled to compare a voltage onthe first capacitor with an error voltage signal representative of anoutput of the PFC converter and to generate the first signal.
 11. ThePFC controller of claim 1 wherein the second integrator includes asecond capacitor coupled to be charged with a difference between ascaled current and a reference current when the power switch is off,wherein the scaled current is representative of the input voltage. 12.The PFC controller of claim 11 wherein a scaling factor of the scaledcurrent is representative of an instantaneous dc input voltage.
 13. ThePFC controller of claim 11 further comprising a second comparatorcoupled to compare a voltage on the second capacitor with an adjustederror signal and to generate the second signal.